Direct conversion programmable power source controller: three-phase input with programmable single-phase output

ABSTRACT

The present invention is a direct conversion programmable power controller that receives polyphase input power that has an input power frequency a polarity. The controller includes a primary chopper for each phase of electrical power. Each primary chopper is electrically connected in wye connection to a transformer. A secondary chopper demodulates and rectifies each phase of power. Each secondary chopper has an input connected electrically to the secondary terminals of the transformer and has an output connected to a load in series connection.

RELATED APPLICATION

This application is a continuation-in-part of the application currentlypending Ser. No. 10/062,054 and filed on Jan. 31, 2002.

FIELD OF THE INVENTION

This invention relates generally to power generation and, morespecifically, to programmable three-phase output voltage and frequency.

BACKGROUND OF THE INVENTION

In most power generation applications, synchronous motors are driven togenerate AC power. When so used, the frequency of AC power output isdependent upon the process used to apply torque to the synchronousmotor. Constant torque at standardized values rarely exists in nature toprecisely turn a synchronous machine. The rotational rate or angularvelocity varies greatly. Because angular velocity is proportionate tothe resulting frequency and voltage of AC power, where the angularvelocity of the torque source is variable, the frequency and voltage ofthe resulting power is variable.

Randomly variable frequency AC power is not very useful. It isimpossible to synchronize such power with a power supply network in acommercially practical manner. Such power cannot drive most applicationsdesigned for 60 Hz. line supply voltages. Few applications can toleratesuch frequency variability.

To overcome the frequency variability in power generation, the solutionshave been of two types, input and output solutions. Input solutions aremechanical and govern the power transfer to the synchronous machine.Output solutions condition electrical power garnered from thesynchronous machine.

Traditionally, design constraints on power processing have requiredselection of components based upon the peak power to flow through thosecomponents rather than to design around the mean as in DC power systems.Periodically recurring peaks in the voltage and current waveforms foreach phase develop recurrent power transfer well above the mean.Nonetheless, the power drawn from such a system as an aggregate isconstant. For instance, the total power drawn from a balancedthree-phase source by a balanced resistive load is constant. That is$\begin{matrix}\begin{matrix}{{P_{t}(t)} = {\left( {V^{2}/R} \right)\left\lbrack {{\sin^{2}\omega \quad t} + {\sin^{2}\left( {{\omega \quad t} + \varphi} \right)} + {\sin^{2}\left( {{\omega \quad t} + {2\varphi}} \right)}} \right\rbrack}} \\{= {1.5\left( {V^{2}/R} \right)}}\end{matrix} & (1)\end{matrix}$

where P_(t)=time value of power

V =peak line voltage

R =load resistance (per phase)

ω=source frequency

φ=2π/3

This fact is exploited in the cycloconverter at discrete frequencies.Small fluctuations in frequency are passed to the output. This need notbe the case. The power transferred from source to load is not a functionof time. The transfer of power can be accomplished without storingenergy within the processor between input voltage cycles.

Because the power is independent of time and therefore phase, thereexists a means, by aggregating the outputs from the phases to subjectseveral of the components of the system to constant rather than varyingpower over time. The traditional input solution to electrical powergeneration in applications such as aircraft has been through theConstant Speed Drive (CSD) coupled to a generator providing, forexample, 115 VAC with three-phase power at a constant 400 Hz. In morerecent times this arrangement has combined the CSD and generator into anIntegrated Drive Unit, or IDU. With a constant frequency power outputthis has been a creditable solution, albeit expensive to buy and tomaintain.

More recently, the output solution has been the Variable Frequency (VF)and cycloconverter systems. Cheaper of these two options, VF presentsthe load with power such as 115 VAC, three-phase power but only has adistribution capability at a frequency proportional to the engine speed.For a turbofan engine, for instance, this is usually 2:1. However,because of the wide range of frequency variation, power conditioningwould be essential for almost all cases and, when added to theprocurement, the additional cost of attendant motor controllers becomesprohibitively expensive.

In recent years, power has been generated with a Variable Speed ConstantFrequency (VSCF) cycloconverter. A cycloconverter is a power electronicsdevice designed to provide a variable voltage, constant frequency ACdrive in a one stage operation suitable for supply to an AC motor. Thesedevices work by generating very high frequency three-phase power thenselectively drawing voltages from the peaks of the three phases in amanner to construct rough approximations of lower frequency waveforms.

While a VSCF system unit (cycloconverter) does have the ability toproduce AC and DC simultaneously, it does not produce clean waveforms.The voltage regulation is accomplished by a series of magneticamplifiers, transformers, and bridge rectifiers. The VSCF drive uses asimple drive system and lets the alternator produce an electrical supplythat is frequency wild, i.e. not well-controlled, which is then shapedby a solid-state electrical unit. Nonetheless, the resulting waveformincludes several harmonics that impart an imaginary component to thepower and may interfere with the function of the load.

Still another means of generating constant frequency power from avariable source of torque is based upon converting and rectifying powerto DC before inverting the DC power to AC power such as 60 Hz. linevoltage. This approach requires a converter that can sink current ofopposite polarity to the output voltage. However, most converters cannotaccommodate a non-unity power factor load. Additionally, for a givenpower level, in a single phase rectifier, the current pulses thatcomprise the ripple may be four to five times as large as the peak ofthe current waveform for an equivalent unity power factor load. Suchcurrent requires much larger conductors to minimize resistive losses. Inpolyphase rectifiers, the current peaks are not as large since smallerpeaks occur more frequently, but the power quality is not as goodbecause of large current transitions

Both the cycloconverter and the devices for converting and invertingpower use, transformer rectifier units that operate at line frequencychop power at low frequencies creating low frequency fundamentalsinusoids thus requiring large and heavy transformers and largecapacitors to store energy for smoothing peaks and filling valleys inthe waveform. Introduction of such elements often adds reactive factorsthat affect the power factor. In such configurations the reactancecauses the current to either “lead” the voltage or to “lag” the voltage.Like the input solution, power conditioning is necessary forpower-factor correction. Often this includes the use of synchronousmotors spinning in “no load” states. All of these solutions prove to becostly. The need for constant frequency power has justified thesesolutions.

There exists, then, an unmet need for a power unit for convertingpolyphase variable frequency AC power to constant frequency AC polyphasepower while maintaining a unity power factor. Internal control of themagnitude of the voltage would further enhance the utility of such apower unit.

SUMMARY OF THE INVENTION

The present invention is a direct conversion programmable powercontroller that receives polyphase input power that has an input powerfrequency a polarity. The controller includes a primary chopper for eachphase of electrical power. Each primary chopper is electricallyconnected in wye connection to a transformer. A secondary chopperdemodulates and rectifies each phase of power. Each secondary chopperhas an input connected electrically to the secondary terminals of thetransformer and has an output connected to a load in series connection.

The inventive device exploits high frequency modulation of the signal tobalance the power throughout the input voltage cycle. By modulating boththe width and the amplitude of the power waveform, no power is lost andthe power factor remains substantially the same as that of the load whenthe power factor is calculated at the input frequency.

The invention receives polyphase power of variable frequency and passespower of a set frequency. Relationships between phases are exploitedaccording to known trigonometric identities. Each phase of the power ismodulated distinctly until at the terminals of the transformer, wherethe output voltage represents the difference in potential pairs ofterminals. To describe the invention here, the circuitry for handling asingle phase of the power passed by the inventive device, For thatpurpose, an exemplary phase of the polyphase power supply is set forthhere.

The present invention comprises a system and method for a directconversion programmable power source controller. The invention controlsthree-phase power by accepting a reference signal; generating threemodulating signals, each for one phase of the input power and having afrequency equal to the sum of the input frequency and the desired outputfrequency; accepting the three-phase electrical power from a wyeconnected power source; creating a high, chopping frequency forsampling; phase-angle-modulating the source signal at the choppingfrequency according to each of the modulation signals. While in the highfrequency state, the power passes easily through small two-windingtransformers used either for isolation or for stepping up or steppingdown the power. Once through the transformer, the invention thenre-reverses the reversed power and chops a portion of each half-cycle ina manner that restores the pulse-width modulation of the power sinusoidby the modulating sinusoid. Reversing the polarity of each phase ofelectrical power at the phase-angle modulation frequency to reproducethe pulse-width modulated waveform at the output.

Filters on the output integrate the power waves before feeding the powerto the load in delta. Between any two terminals of the delta connection,the output voltage presents as a well-formed sinusoid.

According to further aspects of the invention, the invention convertsthree-phase AC power to programmable frequency three-phase power withoutan intermediate DC stage and without large power storage devices.

The invention presents the source with the same power factor as theload.

According to the invention, conversion to high frequency reduces thesize of necessary isolation transformers.

Also, inclusion of small high-frequency transformers allows correctionfor low-line conditions.

Further, the high-frequency conversion technique reduces the size of theinput and output filters and allows those filters to produce nearperfect sine waves.

Users can also optionally program output voltage and frequencies withoutcomponent change by providing appropriate reference waveforms.

The invention also allows for bi-directional power flow thus toaccommodate non-unity power factor (reactive), or braking orregenerative loads.

Wide separation of conversion and output frequencies eliminates phaseerrors due to input and output filters. The presence of a phasereference allows comparison with output waveforms for control loops. Thecontrol loops eliminate line disturbances such as phase-voltageimbalance and short-term spikes.

Further, the control process does not require digital signal processing.The inventive device may operate with a state machine with a look-uptable for the desired waveforms, and a standard multiplyingdigital-to-analog converter.

The present invention provides means of converting three-phase power ofvariable frequency and amplitude to three-phase power of programmableand constant frequency and amplitude without an intermediate DCconversion.

BRIEF DESCRIPTION OF THE DRAWINGS

The preferred and alternative embodiments of the present invention aredescribed in detail below with reference to the following drawings.

FIG. 1a is a schematic diagram of a single phase of the inventive powercontroller;

FIG. 1b is a block diagram placing the single phase as shown in FIG. 1ain context of a three-phase power supply;

FIG. 2 is chart comparing the two constituent logic waveforms and therepresentative output of the two waveforms through an XOR gate;

FIG. 3 is a detail schematic of the gating circuit used in the primaryand secondary choppers;

FIG. 4 is the tracing of a power wave pulse-width modulated by a secondsine wave;

FIG. 5 is detail schematic of the gate driver circuit used in thesecondary chopper;

FIG. 6 is the tracing of the three resultant sinusoids from each of thethree phases; and

FIG. 7 is a flowchart setting forth the method of presenting three-phaseprogrammable frequency and voltage to a load.

DETAILED DESCRIPTION OF THE INVENTION

The inventive device exploits high frequency modulation of the signal tobalance the power throughout the input voltage cycle. By modulating boththe width and the amplitude of the power waveform, no power is lost.Referring to FIGS. 1-7, the present invention is a direct conversionprogrammable power controller that receives polyphase input power thathas an input power frequency a polarity. The controller includes aprimary chopper for each phase of electrical power. Each primary chopperis electrically connected in wye connection to a transformer. Asecondary chopper demodulates and rectifies each phase of power. Eachsecondary chopper has an input connected electrically to the secondaryterminals of the transformer and has an output connected to a load inseries connection.

The invention receives polyphase power of variable frequency and passespower of a selectable frequency. Load current reflected to the source issimilarly modulated so that the source current frequency tracks changesin the source voltage. Relationships between phases are exploitedaccording to known trigonometric identities. Each phase of the power ismodulated distinctly until at the terminals of the transformer where thevoltage represents the difference in potential at distinct terminals ofthe transformer. To describe the invention here, FIG. 1a portrays thecircuitry for handling a single phase of the power passed by theinventive device. For that purpose, an exemplary phase of the polyphasepower supply is set forth here.

Referring to FIGS. 1a and 1 b. A generator 20 provides one of severalphases of power to the circuit 10 shown as is shown in FIG. 1b. In mostcases, this will be a driven three-phase synchronous machine connectedin wye. The generator 20 supplies a voltage) to a pair of primaryterminals of a transformer 55. A secondary pair of terminals of thetransformer 55 then feeds power to or receives it from a load 80. Theinvention will receive current from a load as in the case ofregenerative braking of a load.

The inventive circuit includes inductive and capacitive filter elementson the primary circuit, 42 and 46 respectively, and inductive andcapacitive filter elements on the secondary circuit, 72 and 76respectively. The filters, 42 and 46, are used to remove ripple from thesource currents and load voltages. Because the filters need only removevery high frequency components of the signals, the filter elements 42and 46, 72 and 76 need not store significant power. Finally, inelectrical connection to the primary and secondary terminals of thetransformer 55 there are a primary chopper 50 and a secondary chopper60.

FIG. 1a is a schematic diagram of a circuit corresponding to a singlephase of the inventive system. The system, in fact, comprises multiplecircuits of the configuration shown in FIG. 1b. For purposes ofillustration, this exposition will present a three-phase system. Thoseskilled in the electrical arts will readily see that the same inventivesystem is adaptable to any multiple-phase execution by replicating thecircuit shown in FIG. 1a to correspond with each phase.

The primary chopper 50 performs one of the main roles, that of phasemodulating the power signal in the inventive power supply, by deriving amodulation sinusoid in a feedback loop as set forth below. Thismodulation sinusoid as derived from the source voltage and referencesignals, has a signal logic voltage level and a frequency representingthe sum of the frequencies of sinusoids from the input power sourcesinusoid and a reference sinusoid at the desired output frequency. Thisreference sinusoid is, if desired, drawn from a power grid the powersource 20 will supply.

The basic relationship between the elements of the circuit 10 exploitsthe process of modulating the incoming signal with a local oscillatorfrequency. The modulation is the multiplication of the power waveform bythe reference sinusoid. This modulation suppresses the two fundamentalfrequencies as well as produces sum and difference frequencies. Themodulation of the power signal relies upon a trigonometric identityexpressed as follows: $\begin{matrix}{{\sin \quad u \times \sin \quad v} = {\frac{1}{2}\left\lbrack {{\cos \left( {u - v} \right)} - {\cos \left( {u + v} \right)}} \right\rbrack}} & (2)\end{matrix}$

Thus the arithmetic description of the modulation of the first phase ofthree-phase power from a synchronous generator might look like this:$\begin{matrix}{{V_{A} = {{\Phi_{A} \times M_{0}} = {{V_{peak}\left\lbrack {\sin \left( {\omega_{s}t} \right)} \right\rbrack} \times {\sin \left( {\omega_{m}t} \right)}}}}{V_{A} = {\frac{V_{peak}}{2}\left( {\cos \left( {{t\left( {\omega_{s} - \omega_{m}} \right)} - {\cos \left( {t\left( {\omega_{s} + \omega_{m}} \right)} \right)}} \right)} \right.}}} & (3)\end{matrix}$

where V=peak phase voltage

α=ω_(s)t

β=ω_(m)t;

The ω_(s) and ω_(m) are the source and the modulating frequenciesrespectively

In a three-phase power system, each phase is offset from the otherphases by 2π/3 radians, or 120 degrees. The selection of the phase ofthe modulating frequency yields algebraic opportunities to suppress theupper resulting sideband from the modulation. Where the phase of thepower is offset by 2π/3 radians the modulating frequency is offset by−2π/3 radians or its phase equivalent 4π/3 radians. If the phases aredesignated A, B, and C, the equations describing each of the remainingindividual phases are: $\begin{matrix}{{{V_{B} = {{\Phi_{B} \times M_{-}} = {{V_{peak}\left\lbrack {\sin \left( {{\omega_{s}t} + \frac{2\pi}{3}} \right)} \right\rbrack} \times {\sin \left( {{\omega_{m}t} - \frac{2\pi}{3}} \right)}}}}{V_{B} = {\frac{V_{peak}}{2}\left( {{\cos \left( {t\left( {\omega_{s} - \omega_{m} + \frac{4\pi}{3}} \right)} \right)} - {\cos \left( {t\left( {\omega_{s} + \omega_{m}} \right)} \right)}} \right)}}}{V_{B} = {\frac{V_{peak}}{2}\left( {{\cos \left( {t\left( {\omega_{s} - \omega_{m} - \frac{2\pi}{3}} \right)} \right)} - {\cos \left( {t\left( {\omega_{s} + \omega_{m}} \right)} \right)}} \right)}}} & (4) \\{V_{C} = {\frac{V_{peak}}{2}\left( {{\cos \left( {t\left( {\omega_{s} - \omega_{m} + \frac{2\pi}{3}} \right)} \right)} - {\cos \left( {t\left( {\omega_{s} + \omega_{m}} \right)} \right)}} \right)}} & (5)\end{matrix}$

The algebra now reveals the presence of a cosine sum term,cos(t(ω_(s)+ω_(m))), present in each modulated phase. The presence ofthis cosine sum term in each phase presents an opportunity to eliminateit by subtracting one phase from another. Connecting the three outputsin delta such that the voltage across any pair of terminals yields amathematical expression of the difference between any two terminals asoutput expressions: $\begin{matrix}{V_{A - B} = {V_{peak}\frac{\sqrt{3}}{2}\left( {\sin \quad {t\left( {\omega_{s} - \omega_{m} + \frac{2\pi}{3}} \right)}} \right)}} & (6) \\{V_{B - C} = {V_{peak}\frac{\sqrt{3}}{2}\left( {\sin \quad {t\left( {\omega_{s} - \omega_{m}} \right)}} \right)}} & (7) \\{V_{C - A} = {V_{peak}\frac{\sqrt{3}}{2}\left( {\sin \quad {t\left( {\omega_{s} - \omega_{m} - \frac{2\pi}{3}} \right)}} \right)}} & (8)\end{matrix}$

Each of the arithmetic processes set forth above can be effected in thecombination of the primary chopper 50, the transformer 55, and thesecondary chopper without the use of digital logic at power levels.“Chopping” occurs at a frequency significantly higher than the frequencyof either the power supply or the desired output. According to theinvention, the multiplication is a byproduct of the chopping process inconjunction with pulse-width modulation. Pulse-width modulationcharacterizes the output of the secondary chopper 60, that is themultiplication modulation. The width of each pulse represents the valuederived from the sinusoidal modulation function, the height of thepulse, from the power sinusoid. Thus, when the resulting waveform isintegrated over time, the resulting waveform is a smoothproduct-of-sines.

The pulse-width modulation is implemented by means of two distinctsteps. First, the primary chopper 50 achieves phase-angle modulation.Then, once so modulated, the circuit 10 passes near-square-waves throughthe transformer 55 at chopping frequency that is very high relative tothe power and reference sinusoids. The synchronous secondary chopper 60inverts the second of each square wave pair to create apulse-width-modulated waveform at the same chopping frequency.

The ability to create a phase-angle modulated square wave arises fromparameters implicit in the nature of a sine wave itself. The magnitudeof the sine function is bounded by the value one or unity, i.e.:

|sin x|≦1, for all x  (9)

Without resorting to digital logic and without amplification, thisrelationship allows the modulation of the power waveform solely withswitching devices. Rather than to mathematically multiply theinstantaneous values of the power sinusoid with the instantaneous valuesof the reference sinusoid at each given moment, t, to produce a smoothfunction, the invention admits power during a portion of a samplingperiod corresponding to the value of the reference sinusoid.

Before considering the modulation of a power waveform by means of theprimary chopper 50, consider the chopping of a simple constant DCvoltage. “Chopping,” as used, means the opening and closing of aswitching device at a given frequency. A full chopping cycle comprisesone period when the switch is in an open or non-conducting state and animmediately adjacent period when the switch is in a closed or conductingstate. In the context of chopping, the duty cycle is the proportion theperiod of the conductive state bears to the whole cycle. Chopping isoften used to supply a lesser DC voltage to a load from a higher DCvoltage. For each segment, the power is chopped according to thefollowing equation:

V _(out) =V _(in) ×D, where D is the duty cycle  (10)

To modulate the power wave according to a second sine wave, one can varythe duty cycle according to the magnitude of the sine. Thus, for eachperiod of one sampling interval, Δt, the switch will remain closed (orconductive) for a smaller period in proportion to the value ofsin(ω_(m)nΔt). $\begin{matrix}{{{For}\quad {each}\quad {integer}\quad n},{{{and}\quad {for}\quad \Delta \quad t} = \frac{1}{\omega_{samp}}},{V_{out} = {V_{in} \times {\sin \left( {\omega_{m}n\quad \Delta \quad t} \right)}}}} & (11)\end{matrix}$

Outside of the bounds of the first lobe of the sine wave, the value ofthe sine function goes negative. Equation 11 would normally be veryproblematic when seeking to modulate the power sinusoid. A duty cyclecannot become negative in length where the sine term becomes negative.It is for this reason that an intermediate step makes sense.

This average output is the key to the invention's two-step modulation.The primary chopper modulates the power signal by reversing the polarityat a high frequency. The onset of the reversal occurs after a fixedclock signal according to the term 0.5×(1+M sin(ω_(m)nΔt)). Thus for aphase angle Φ, the phase delay of the chopped power wave is:$\begin{matrix}{{\Phi \left( {n\quad \Delta \quad t} \right)} = {0.5\left( {1 + {M\quad {\sin \left( {\omega_{m}n\quad \Delta \quad t} \right)}}} \right)\frac{2\pi}{\Delta \quad t}}} & (12)\end{matrix}$

This primary chopping allows a balanced high-frequency power wave topass through the necessary transformer 55. After the power passesthrough the transformer 55, the secondary chopper re-reverses thepolarity of the power wave according to the fixed frequency clocksignal, folding the power signal onto itself according to the constantterm in the expression 0.5×(1+M sin(ω_(m)nΔt)). Because the bridgeoperation inverts the polarity of the incoming voltage, the average ofthe bridge output is zero at D=0.5. As D varies from zero to 1, theoutput of the bridge varies from 100% negative polarity to 100% positivepolarity; or from −1 to 1 times the source voltage. The synchronized andperiodic reversal of the polarity by the secondary chopper suppressesthe power wave causing the averaged output of the bridge to be:

Vout=0+M sin(ω_(m) nΔt)  (13)

The M term is a coefficient between zero and one, chosen for purposes ofscaling the modulation, increasing or decreasing the magnitude of thesine curve corresponding to regulating the voltage output in theequation (10).

One of the virtues of the inventive primary chopper 50 is presentationof phase-angle-modulated square waves to the primary terminals of thetransformer 55. Rather than presenting a pulse-width-modulated squarewave of the power at the sampling frequency, the invention seeks only toaccomplish half of the process of multiplying the sinusoids beforeplacing the power through a transformer 55 (isolation, “step up,” or“step down” as the determined by the application). Rather than tomodulate the width of the pulses according to the sine, the voltage sentthrough the transformer as represented in Equation 13 is a phasemodulated square wave.

The need for using a nearly true square-wave over apulse-width-modulated wave is that the pulse-width-modulated waveformcontains very low frequency components, much lower than those of thechopper square wave. To achieve this primary chopping of the waveforms,a frequency significantly higher than either the power or the modulatingsine waves is selected. This is the frequency that will be passedthrough the transformer and is suitably any frequency selected tominimize hysteresis and eddy current losses while remaining high enoughto yield good resolution of the power wave. Frequency might be chosen tooptimize the characteristics of a chosen transformer. So long as it issuitably high to yield sufficiently formed sinusoids when passed throughthe final filters, the frequency is not critical to the operation of theinvention.

A square wave that is phase modulated contains all of the informationnecessary to effect the product of sines modulation, but the fundamentalfrequency will be very close to the frequency of the phase modulatedsquare wave. This fact yields an opportunity to select a frequency muchhigher than that of the input power wave and the modulation sine wave.Doing so will allow the use of a much smaller transformer in theinvention as transformers designed for high frequency are generally muchsmaller than those necessary for lower frequency so long as the averagevoltage sent through the transformer is zero. The higher the lowestfrequency, the shorter the period before the average returns to zero.

Transformers also require symmetrical waves for efficiency. Where theaverage voltage across the primary terminals over a period exceeds zero,the net resultant current will push the transformer towards coresaturation. A symmetrical (the voltage peaks displace equally from zero,i.e. the magnitude of the negative and positive portions of the waveformis the same) will fulfill the requirement that the average is zero overtwo sampling periods. The higher frequency square wave allows thetransformer to pass power without a long-term (greater than a singleperiod at the higher frequency) effect on the transformer flux. Thus itis advantageous to pass symmetric square waves through the transformerand only then to further chop the phase modulated square waves tore-create the “product of sines” pulse-width-modulated waves discussedabove.

In FIG. 2, the logic signals for effecting Equation 12 are shown. Curve110 is a clock curve used to set the sampling frequency, ω_(samp), forpurposes of admitting a portion of the power waveform. The curve 110 hasa period of Δt, reflecting a sampling period. ωt is used here to preventconfusion with the period of either the power source or modulatingsinusoids and to reflect the analogy to Riemann Sums. The next portionof the algorithm necessary to effect Equation 12 is the curve relatingto the 0.5(1+M sin(ω_(m)nΔt)) factor. To effect that portion of theequation, the triggering of gates in the primary chopper 50, discussedbelow in FIG. 3, will delay the onset of a square-wave 120 with a phaseangle delay of Φ(nΔt)=0.5(1+M sin(ω_(m)nΔt))2π/Δt radians where ω_(m) isthe frequency of the modulating sinusoid 130 referred to above. For eachperiod, Δt, commencing at either the leading or the falling edge of theclock pulse 122, the pulse will remain logically high for a proportionof the Δt period equal to 0.5(1+M sin(ω_(m)nΔt)) and then immediatelyfall to negative logic levels 124. Sending the clock curve 110 and thepulse-width-modulated square-wave pulse through an XOR gate results inthe curve shown as curve 140. The result is a digital gate capable ofdirecting switching at a rate to effect the 0.5(1+M sin(ω_(m)nΔt)) phasemodulation.

Curve 140 is a phase-angle-modulated square-wave with glitches 142. Indigital logic, a glitch is a sudden break in function or continuity,usually caused by switching error, of a transient nature. Here, duringthe period when curves 110 and 120 simultaneously transition from low tohigh or high to low, a glitch occurs. Because the logic is keyed to theclock curve 110, this glitch 142 is very transient but, it is,nonetheless, noted here. The glitch has no functional significance inthis application because the inductive properties of the transformerremove the affects of the glitch 142 from practical consideration.Nonetheless, the glitch 142 serves as a benchmark for discussion oflater processing.

With the resulting phase-angle-modulated curve 140, the invention iscapable of appropriately chopping the incoming power curve. Becausepower line voltage levels would burn up most digital logic chips, anappropriate gating apparatus is necessary. FIG. 3 portrays a schematicdiagram of one an exemplary gating device 52 that is a part of theprimary chopper 50. Although any suitable switching device will serve,the, gating device 52 comprises gangs of MOSFETs 501, 502, 503, and 504;that is the gating device 52 is a simple bridge. A logic signal fed todiagonal pairs of MOSFETs (e.g. 501 and 503 or 502 and 504) opens andcloses the MOSFETs to appropriately pass the power curve to thetransformer 55.

As indicated in the discussion of FIG. 2, curve 140 is the output of anXOR gate (not shown). Most commercial XOR gates also provide thecomplementary logic curve. In the complement, logical highs in theoriginal curve correspond to logical lows in the complement and viceversa. The output of the XOR gate is fed to the gate terminals 15 and 17of one diagonal pair of MOSFETs 501 and 503, respectively. Thecomplementary output is fed to the gate terminals 16 and 18 of theremaining diagonal pair of MOSFETs 502 and 504, respectively. By thislogical scheme, a voltage potential corresponding to the amplitude ofthe power curve is fed to the primary side of the transformer 55, withrapidly reversing polarity according to the logical transitions of theoutput of the XOR gate. The waveform at the primary terminals at thetransformer 55 is a square wave at the clock frequency and power levels;it is offset from the clock wave by a time period corresponding to themagnitude of the modulating sine according to Equation 13. The wave ishigh frequency and symmetrical and thus allows the use of a smallertransformer.

After the primary chopper 50 sends this output wave through thetransformer 55, the secondary chopper 60 demodulates the waveform so asto remove the half amplitude power wave that remains:

 V _(out)(nΔt)=V _(in)×[0.5(1+M sin(ω_(m) n2πΔt))]−0.5V _(in) =MV _(in)sin(ω_(m) n2πΔt)  (14)

The secondary chopper 60 uses one half-cycle of the clock pulse 110 toinvert that portion of the wave that either precedes or follows theclock transition 2. By this means, the secondary chopper inverts thecurve between every other glitch portrayed in FIG. 2. The same gatingdevice 52 shown in FIG. 3 is suitably used in the secondary chopper. Theresulting pulse-width-modulated curve 120 now determines the periods ofconductivity, and the clock transition 110 causes the reversal of thepolarity of the output. Because the demodulation is synchronized to themodulation, the resulting waveform is the product of sines of Equation(3) shown in FIG. 4 as curve 160. By conducting that output to the loadin delta and filtering the same with a low pass LC filter, the inventionproduces the power described in Equations 6, 7, and 8: smooththree-phase power at the desired frequency.

Another inventive aspect of the present invention is the use of afeedback loop to produce a waveform as the sum of frequencies, ω_(m),used for modulation. As indicated in Equation 3 above, the modulatingfrequency, ω_(m), will have a value of the sum of the desired outputfrequency and the source frequency. As frequencies are not readilyadded, the trigonometric identity set forth in Equation 2 is, again,employed to generate the modulating frequency, ω_(m). The interrelationof the power and control circuits provides a feedback loop that assuresthe regularity of the output waveform by assuring that the differencebetween the input frequency and the sum of the input and modulatingfrequencies remains the same.

To maintain this difference in the feedback loop, the invention drawsfrom the input power curves from each phase of the power from the source20. For each phase, the inventive device uses as input a referencewaveform representing the desired output frequency. This might be awaveform drawn from the power grid, from a local oscillator or from alook-up table. Using this source frequency, ω_(s), from above and(ω_(ref), the invention generates a product of the sines according tothe same trigonometric identity: $\begin{matrix}{\Gamma_{A} = {{\sin \quad \omega_{s}t \times \sin \quad \omega_{ref}t} = {\frac{1}{2}\left( {{\cos \left( {t\left( {\omega_{s} - \omega_{ref}} \right)} \right)} - {\cos \left( {t\left( {\omega_{s} + \omega_{ref}} \right)} \right)}} \right)}}} & (15) \\\begin{matrix}{\Gamma_{B} = {{\sin \left( {t\left( {\omega_{s} + \frac{2\pi}{3}} \right)} \right)} \times {\sin \left( {t\left( {\omega_{ref} + \frac{2\pi}{3}} \right)} \right)}}} \\{= {\frac{1}{2}\left( {{\cos \left( {t\left( {\omega_{s} - \omega_{ref}} \right)} \right)} - {\cos \left( {t\left( {\omega_{s} + \omega_{ref} - \frac{2\pi}{3}} \right)} \right)}} \right)}}\end{matrix} & (16) \\\begin{matrix}{\Gamma_{C} = {{\sin \left( {t\left( {\omega_{s} - \frac{2\pi}{3}} \right)} \right)} \times {\sin \left( {t\left( {\omega_{ref} - \frac{2\pi}{3}} \right)} \right)}}} \\{= {\frac{1}{2}\left( {{\cos \left( {t\left( {\omega_{s} - \omega_{ref}} \right)} \right)} - {\cos \left( {t\left( {\omega_{s} + \omega_{ref} + \frac{2\pi}{3}} \right)} \right)}} \right)}}\end{matrix} & (17)\end{matrix}$

The mathematics is easily performed at signal levels using an analogfour-quadrant multiplier, a multiplying digital-to-analog converter, orstate machine recalling a sine form from a memory look-up table, forexample. The feedback loop employs one small but critical differencebetween feedback and the forward algorithms is the direction ofrotation, or phase sequence, of the reference sine multipliers. For thefeedback path, the direction is the reverse of the modulator set shownin the forward path. That is why the sum frequency, rather than thedifference frequency, is preserved. The difference between phasesresults in a simple sine wave at a frequency equal to the sum of theinput and modulation frequencies. For instance, as between the B and Cphases: $\begin{matrix}{{{\Gamma_{B} - \Gamma_{C}} = {\frac{1}{2}\left( {{\cos \left( {t\left( {\omega_{s} + \omega_{ref} + \frac{2\pi}{3}} \right)} \right)} - {\cos \left( {t\left( {\omega_{s} + \omega_{ref} - \frac{2\pi}{3}} \right)} \right)}} \right)}}{{\Gamma_{B} - \Gamma_{C}} = {\frac{1}{2}\left( {2\left( {\sin \quad {t\left( {\omega_{s} + \omega_{ref}} \right)}} \right)\left( {\sin \left( \frac{2\pi}{3} \right)} \right)} \right)}}{{\Gamma_{B} - \Gamma_{C}} = {\left( {\sin \quad {t\left( {\omega_{s} + \omega_{ref}} \right)}} \right)\left( \frac{\sqrt{3}}{2} \right)}}} & (18)\end{matrix}$

The result is a sinusoid at a frequency representing the sum of theinput and the reference frequencies, ω_(m), with the modulator for theinput power A phase. The other two possible differences will yieldsimilar results for the B and C phases of the input. Where the sourcefrequency departs from a given value, the difference of frequencies termin Equations 3, 4, and 5 will remain constant; the sum of frequenciesterms cancels in the resulting Equations 7, 8, and 9 and therefore hasno effect upon the resulting waveform. The inventive device reliablyprovides constant frequency power.

With the feedback process in place, the invention is self-regulatingsuch that fluctuations in the input frequency do not affect the outputfrequency. So, too, as earlier identified, the amplitude of the outputsine is governed by the value of the modulating constant, M, in Equation14. M can be used to control a simple feedback loop. A distinctpreferred embodiment allows the gating device 52 shown in FIG. 3 to besuitably used in a passive embodiment of the secondary chopper. Ratherthan triggering gates 15, 16, 17, and 18, a reference voltage can besent to the gates allowing the gates to rectify the output of thetransformer 55 (FIG. 1) and in that way determine the periods ofconductivity. In this manner, purely passive gating will demodulate toDirect Current without any secondary chopper drivers in the passivegating embodiment.

It is worthwhile to touch on the resulting unity or near unity powerfactor. Recalling that at the stage of the inventive device where thepower is routed through the transformer 55, the square-waves are at afrequency much higher than that of either the source or the outputpower. After secondary chopping at the chopper 60, the fundamentalfrequency does drop but still remains much higher than the source andinput power frequencies. The inductive filter element 72 (FIG. 1a) andthe capacitive filter element 76 (FIG. 1a) can energize and de-energizeat frequencies that are two orders of magnitude higher than that of theoutput power. It will be appreciated, though, that the filter elements72 and 76 may energize and de-energize at frequencies that are eithergreater than or less than two orders of magnitude higher than that ofthe output power, as desired. Thus such leading and lagging effects thatthey might introduce are de minimus at the output.

FIG. 5 depicts a secondary chopper driver for the invention. Thesecondary chopper has two purposes: to periodically reverse the polarityof the output of the single phase according to the clock curve 110; andto reverse the polarity additionally anytime the voltage difference ofthe filtered output of the phase should go negative thereby folding thevoltage to maintain the positive voltage across the output terminals ofthe secondary chopper section 75. As indicated in the discussion of themodulating constant M above, the inventive device must constantlymonitor the output voltage and modulate the output of each branchaccordingly to keep the output of the branches in balance. Themathematic relationships disclosed in the equations above rely upon therelative voltage levels remaining constant.

In a preferred embodiment of the secondary chopper driver 301 monitorsthe voltage at terminals 305 and 306 in FIG. 1a. Monitoring voltageacross terminals 305 and 306 is preferred because these terminalspresent a voltage that is filtered to most closely approximate thesinusoids of ideal alternating current. In this preferred embodiment,the instantaneous voltage across the terminals 305 and 306 reveals themoment appropriate for reversing the polarity to maintain a constantlypositive voltage across the terminals.

The analog-to-digital converter 310 presents a digital signalrepresentative of the instantaneous voltage across the terminals 305 and306. The signal will repeatedly change according to a clock signaldriving the analog-to-digital converter 310. Acting to time theinstantaneous sampling by the analog-to-digital converter 310, the clockpresents data points along the output sinusoid.

In a preferred embodiment of the secondary chopper 301, theanalog-to-digital converter 310 will represent the full range, bothpositive and negative, of the voltages measurable by theanalog-to-digital converter 310. Thus, the negative extreme of range isrepresented by logical lows on all of the output levels; the positiveextreme of range is represented by all logical highs. In such aconfiguration, the most significant bit when high represents a positivenumber, when low a negative. The output of the most significant bit isthen fed to a JK Flip-Flop with a negative input 320 acting as a latch.When so configured, the most significant bit will alternate the outputof the latch with each digital representation of a negative voltage atanalog-to-digital converter 310.

The output of the JK Flip-Flop 320 is fed to one of two inputs on an XORgate 340. The remaining input receives the same curve 110 at 330 thatdrives the primary chopper 50. So configured the XOR gate 340periodically reverses the polarity of the output of the secondarychopper 60 by simultaneously closing switches 501 and 503 as it opensswitches 502 and 504 and then reversing the process. Operation of theXOR gate 340 is best understood by examining the operation as a resultof each input.

The XOR gate 340 is the trigger for the bridge as set out in FIG. 3.Where the voltage level of one input of the XOR gate 340 remainsconstant, the oscillations shown as curve 110 and fed to remaining gatetime the periodic reversals of the logical voltage signals present atoutput and negative output of the gate 340. Thus, as in the primarychopper, the logical signals will trigger paired switches 501 with 503and 502 with 504 periodically, resulting in the periodic reversal ofvoltage across the load. Where the output of the JK Latch 320 is fed tothe other input of the XOR gate 340 such that one input is the 110 curveand the other the output of the JK Latch 320, an additional reversaloccurs such that the filtered output at terminals 305 and 306 willremain positive.

FIG. 6 demonstrates graphically the output of the single phase at theterminals 305 and 306. If the input of the XOR gate 340 representing theoutput of the JK Latch 320 is held constant, the voltage at 305 and 306will follow the curve that is the composite of 412 and 402, a sinusoid.With the oscillations of the JK Latch 320, the voltage does not gonegative and rather follows the curve set out at 421. Over time, thereversals will well approximate the repeated positive lobes of thesinusoid thus alternating between 412 and 421. Across the three phasesshown in FIG. 1b, the outputs will appear as curves 412, 415, and 418respectively. As the output of the three phases is summed, the curve 445results. Selecting appropriately the values of the output filters 72 and76 will remove the ripple shown to result in DC voltage as the summedoutput.

FIG. 7 sets forth a method 205 of producing programmable three-phasevoltage and frequency according to the invention. Electrical power isreceived from a driven synchronous machine at block 210. To constructthe feedback leg, at block 215, each phase of the source power issampled. A reference sinusoid is drawn from either the power grid or agenerated oscillation at block 220.

Having the two distinct logic-level voltage sine waves, the device thenmultiplies the sine values to create a product of sines for each phaseaccording to Equations 15, 16, and 17 at block 225. At block 230,subtracting each phase from the next adjacent phase to result in threedistinct sinusoids, each at a frequency representing the sum of thesource frequency and the desired output frequency, according to Equation18, results in three modulation sinusoids, each for one source phase.

At block 235, the invention employs one of two modes of the feedbackprocess. Because the magnitude of the output voltage is proportionate tothe coefficient used to scale the modulation sinusoid, the methodmonitors the output voltage and adjusts the coefficient between zero andone to achieve a constant or a programmable output voltage. The scalingis a dynamic and constant process thus overcoming fluctuations in thesource voltage. As properly scaled, the modulating sinusoid thendictates the phase-angle modulation of the source power as it travelsthrough the transformer.

The invention phase-angle creates and then continuously shifts asignificantly higher frequency square-wave at block 235. The shifting isaccording to the corresponding modulation sinusoid for the phase of thesource power to be modulated. At block 240, the polarity of thecorresponding source power phase is then periodically reversed accordingto this phase-angle shifted square wave. The resulting power wave issent through the primary terminals of a transformer 55 (FIG. 1a) atblock 245. At block 250, on the secondary side, the resulting voltage ischopped and re-reversed according to the original square-wave to producea pulse-width-modulated product of the source power phase and the “sumof the frequencies” modulating sine wave 160. Filtering the sine tointegrate the resulting wave results in a smooth wave at 255. The poweris supplied to the load in delta connection at block 260.

While the preferred embodiment of the invention has been illustrated anddescribed, as noted above, many changes can be made without departingfrom the spirit and scope of the invention. For example, four, six, oreight phase power is equally susceptible to frequency and voltageprogramming according to the same modulation scheme. Furthermore, manysimplifications to the presented embodiment are possible, includingoperation without a transformer, and the algorithm shown is applicable,or adaptable. Accordingly, the scope of the invention is not limited bythe disclosure of the preferred embodiment. Instead, the inventionshould be determined entirely by reference to the claims that follow.

What is claimed is:
 1. A direct conversion programmable power controllerreceiving polyphase input power at an input power frequency and having apolarity, the controller comprising: a primary chopper for each phase ofelectrical power, each primary chopper having an input, electricallyconnected in wye connection to each phase of power, having an output; atwo-winding transformer for each phase of power, each transformer havingprimary and secondary terminals and connected electrically by itsprimary terminals to the output of the primary chopper; and a secondarychopper for demodulating and rectifying each phase of power, eachsecondary chopper having an input connected electrically to thesecondary terminals of the transformer and having an output connected toa load in series connection.
 2. The controller of claim 1, wherein theprimary chopper periodically reverses the polarity of each phase ofpower from the power source at a frequency significantly higher than theinput power frequency.
 3. The controller of claim 2, wherein the primarychopper periodically reverses the polarity of the phase of power fromthe power source wherein an onset of the polarity reversal within aperiod varies according to a sinusoid.
 4. The controller of claim 3,wherein the sinusoid has a frequency equal to a sum of the frequency ofthe power input frequency and a desired frequency of the output.
 5. Thecontroller of claim 4, wherein a feedback loop generates the sinusoidaccording to the power input frequency and a reference sinusoid.
 6. Thecontroller of claim 5, wherein the feedback loop samples the input powerto determine the power input frequency.
 7. The controller of claim 5,wherein the feedback loop samples a power grid to generate the referencesinusoid.
 8. The controller of claim 5, wherein the feedback loopgenerates the reference sinusoid by means of a state machine.
 9. Thecontroller of claim 5, wherein the feedback loop generates the referencesinusoid by means of a local oscillator.
 10. The controller of claim 5,wherein the feedback loop generates the reference inusoid from valuesstored in a look-up table.
 11. The controller of claim 5, wherein thesecondary chopper periodically reverses the polarity of the phase ofpower from the transformer at the frequency significantly higher thanthe input power frequency.
 12. The controller of claim 11, wherein thesecondary chopper conducts the phase of the power according to a dutycycle within each period and where the secondary chopper periodicallyreverses the polarity of the phase of power.
 13. The controller of claim12, wherein the duty-cycle varies according to the sinusoid.
 14. Thecontroller of claim 12, wherein feedback loop samples the power at theload and varies a coefficient between zero and one according to theamplitude of the sampled power.
 15. The controller of claim 14, whereinthe duty cycle varies according to the product of the coefficient andthe sinusoid.
 16. The controller of claim 12, wherein the secondarychopper modulates the input power applied to the load according to thesinusoid.
 17. The controller of claim 4, wherein the secondary choppercomprises rectifiers periodically reversing the output of thetransformer for each phase of power.
 18. The controller of claim 17,wherein the secondary chopper reverses the output of the transformer foreach phase of power according to the polarity of output of thetransformer.
 19. A method for controlling three-phase electrical power,the power having a polarity and a power frequency, the methodcomprising: generating a first electrical signal, having a firstfrequency greater than the power frequency of the three-phase electricalpower; generating three second signals each of the second signals havinga second frequency and being offset from the other second signals byphase-angles of 2π/3 radians; providing the three-phase electrical powerfrom a wye connected power source, each phase of electrical power beingoffset from the other phases of the electrical power by phase-angles of2π/3 radians; phase-angle modulating the first signal according to eachsecond signal to produce a third signal at the first frequency tocorrespond with each phase of the three-phase electrical power;reversing the polarity of each phase of electrical power at the firstfrequency according to the third signal that corresponds with the phaseof electrical power; providing a two-winding transformer having primaryand secondary terminals for each phase of electrical power; applying thephase of electrical power and the reversed phase of electrical poweracross each set of the primary terminals; chopping each phase ofelectrical power induced across each set of secondary terminals topulse-width-modulate each phase of electrical power according to aninstantaneous value of the second signal; rectifying each phase ofelectrical power to produce a generally biased voltage; applying thephases of electrical power from the secondary terminals to series load.20. The method of claim 19, wherein providing three-phase electricalpower includes filtering the power to remove ripple.
 21. The method ofclaim 19, wherein applying the pulse-width-modulated electrical powerincludes filtering to integrate the output.
 22. The method of claim 19,wherein generating a first electrical signal includes generating asquare-wave.
 23. The method of claim 22, wherein generating a firstelectrical signal includes generating a phase-modulated square-wave. 24.A direct conversion programmable power controller receiving polyphaseinput power at an input power frequency and having a polarity, thecontroller comprising: a primary chopper for each phase of electricalpower, each primary chopper having an input, electrically connected inwye connection to each phase of power, having an output; a transformerfor each phase of power, each transformer having primary and secondaryterminals and connected electrically by its primary terminals to theoutput of the primary chopper; and a secondary chopper for each phase ofpower, each secondary chopper having input connected electrically to thesecondary terminals of the transformer and having an output connected toa load in series connection.
 25. The controller of claim 24, wherein therectifying bridge is a diode bridge.
 26. The controller of claim 24,wherein the rectifying bridge is a thyristor bridge.